Waveguide optical modulators are well known in the art and are used in a variety of applications. For high bandwidth application, for example at modulation rates in the 5 GHz to 40 GHz range, such modulators typically are based on electro-optical materials incorporating voltage-controlled waveguides forming a Mach-Zehnder interferometer structure, which enables to convert phase modulation of light propagating in the waveguides into an optical power modulation of a combined light at the output of the Mach-Zehnder structure. Such modulators are conventionally referred to as Mach-Zehnder (MZ) modulators.
One disadvantage of conventional voltage-controlled MZ structures is that its transmission characteristic, i.e. the dependence of an output optical power on the applied voltage, is substantially nonlinear, and is generally in the form of a sinusoid. This non-linear MZ transmission characteristic of the MZ interferometer (MZI) structure results in non-linear distortions of the modulation of the output optical signal in conventional MZ modulators. For example, if the voltage applied to the electrodes of a conventional MZ modulator has a modulation component at a frequency J and the modulator is optically fed with cw light, the output optical power from the modulator may include not only a modulation components at the modulation frequency f, but also modulation components at harmonics of the modulation frequency 2f, 3f, etc, which are referred to as, respectively, the second order distortion, the third order distortion, etc, which result from the appearance of products of the modulation components of different orders in the output signal due to the nonlinearity. The second order distortions are also referred to as CSO (Composite Second Order) distortions or CSO distortion products, while the third order distortions are also referred to as CTB (Composite Triple Beat) distortions or CTB distortion products. These distortion products are problematic for systems that transmit analog signals or multi-level digital signals that are analog in nature, such as Quadrature Amplitude Modulation (QAM) signals of various formats. In applications where the useful signal is in the modulation of the electrical field E of the output optical wave rather than in that of the optical power, such as for the transmission of QAM signals, the 1st and 2nd order distortions may relate to respective harmonics in the modulation of the electrical field E, which is proportional to a square root of the optical power P.
Various approaches to reducing non-linear distortions in MZ modulators have been disclosed in the art. U.S. Pat. No. 5,161,206 discloses an x-cut microwave linearized modulator (LINMOD) 10 using a parallel topology that is illustrated in FIG. 1. The modulator is formed of two inner MZ modulators 5, 6 coupled in parallel between an input optical coupler 2 followed by two Y-junction splitters 4, and three Y-junction combiners 18, 19 at the output. Each of the inner MZ modulators 5, 6 has an RF signal electrode 11, 12 and one ground electrode 15, with another ground electrode being shared between the MZ modulators. Biasing electrodes are provided for biasing the inner MZI's 5, 6 to quadrature operating points, i.e. the half-power point in the transmission curve, which eliminates 2nd order (CSO) frequency components in the received signal.
In operation, an applied electrical RF signal is split into two signals, V_RF1_in and VRF2_in, having the same phase but different amplitudes, which are separately applied to the signal electrodes 11, 12 for driving respective inner MZ modulators 5, 6. The ratio of the amplitudes of the RE signals is given by the parameter ARF according to Equation 1, where ARF is typically a value less than 1.
                              A          RF                =                              V                          RF              ⁢                                                          ⁢              2                                            V                          RF              ⁢                                                          ⁢              1                                                          (        1        )            
The output of the modulator 10 is linearized with respect to 3rd order distortions, i.e. the CTB, by adjusting a coupling ratio of the coupler 2, defined by the fraction of light in the coupler that stays within the same waveguide receiving input light, and/or the RF signal split ratio ARF. An optimum value of the coupler coupling ratio Rcplr for the linearization is given by equation (2):
                              R          cplr                =                              A            RF            3                                1            +                          A              RF              3                                                          (        2        )            
For example, ARF equal to 0.6 implies that Rcplr must equal 0.178. FIG. 2 shows a plot of normalized optical output power vs. normalized RF voltage at the input to the electrodes for the device shown in FIG. 1. In this figure, the vertical coordinate represents the optical output power from the modulator 10 normalized to the total optical output power available from all optical output ports. The horizontal coordinate represents the RF voltage normalized to a voltage Vπ that results in a π phase shift in the inner MZI having the larger RF signal. This plot is also called a transfer curve for the device.
The RF power ratio ARF of the RF signals must be carefully tuned to obtain an optimum suppression of the third order (CTB) frequency components in the output signal. It must be preserved at all frequencies, hence the frequency response as defined by an electrical-optical-electrical (e-o-e) forward transmission coefficient, or forward (voltage) gain S21 of the two inner MZI structures 5, 6 must be matched to within approximately 0.2 dB. Having a single tunable coupler 2 that is tuned to satisfy Equation 2 makes the design robust to variations in γ that may occur due to fabrication variations in the electrode and waveguides structures.
One disadvantage of the parallel LINMOD 10 is the loss of about half of the optical power due to the coupler having a coupling ratio that does not match the split ratio of the y-junction combiner near the output of the device.
FIG. 3 shows a linearized modulator 20, described in U.S. Pat. No. 5,148,503, which is incorporated herein by reference; it can be viewed as a serial version of the linearized modulator 10. Two conventional MZ modulators 21, 22 are cascaded in a serial fashion with a first coupler 24 therebetween, and an output coupler 26. As before, the S21 (e-o-e) frequency responses must be carefully matched to within tenths of dB. The serial LINMOD 20 has unequal splits in the coupler 24 or 26, but does not result in any appreciable amount of lost power, as can be seen in the transfer curve plot shown in FIG. 4. The conventional MZI structures 21, 22 are biased such that the LINMOD 20 outputs are at the quadrature (half-power) point without any RF applied. As with the parallel LINMOD 10, both CSO and CTB distortion products are suppressed if the ratio γ of the modulating voltages for the two modulators is suitably selected.
An optimum value of a ratio between the amplitudes of RF signals applied to the signal electrodes 23 and 24 of the MZ modulators 21 and 22, R=V_RF2_in/V_RF1_in, is given by a real root to Equation 3 hereinbelow. The variables γ1 and γ2 relate to the coupling ratios Rcplr1 and Rcplr2 for the couplers 24 and 25, respectively in FIG. 3, as defined by equations 4 and 5. The coupling ratios are defined as the fraction of light that remains within a waveguide after passing through the coupling region of the coupler.R3+3 cos(2γ1)R2+3R+[sin(2γ1)cot(2γ2)+cos(2γ1)]=0  (3)cos2(γ1)=Rcplr1  (4)cos2(γ2)=Rcplr2  (5)
One major problem with the prior art linearized modulators 10, 20 is the need for two separate, but identical RF electrodes being driven by outputs of an RF splitter. Typically the two RF electrodes are slightly different due to variation in the fabrication process across the device. In addition, separate cables and/or RE traces that route the RF signals to each of the signal electrodes may be slightly different in attenuation or length, causing differences in frequency response and/or phase response due to timing skew. Any RF reflections that are not exactly matched will also result in ripple in the frequency response that is different for the two RF electrodes. In practice, these difficulties can be overcome for modulation frequencies of less than 5 GHz, but become very difficult at frequencies approaching or above 20 GHz.
FIG. 5 shows another modulator topology for suppressing CTB frequency components described in G. E. Betts, “A linearized modulator for high performance bandpass optical analog links,” IEEE Microwave Symposium Digest, Vol. 2, May 23-27, 1994, pp. 1097-1100, which is incorporated herein by reference. The LINMOD 30 illustrated in FIG. 5 may be referred to as a carrier suppressed modulator, as the DC component of light is suppressed at the modulator output. In this modulator, a single MZI structure 33 with single RF electrode 31 is nested within a larger MZI structure 35. A coupler 36 can be either at the input or output of the LINMOD 30, or both. DC bias voltages are applied to bias electrodes 37, 38 shown in the drawing. FIG. 6 shows the transfer curve for the modulator shown in FIG. 5. Unlike other LINMODs 10 and 20 described hereinabove, this device is not biased to the half power point. The outer arm 39 of the outer MZI 35 is biased so that both arms of the outer MZI 35 have the same phase or 180 degrees out of phase, when the inner MZI 33 is fully on, i.e. is at a point of maximum transmission. The inner MZI 33 is biased at the point where the waveguide arms thereof have about 0.9 π phase difference between them. This bias point is shown in FIG. 6 indicated by an arrow. The DC component of light at the output of the device is only about 8% of the input power for an idealized lossless modulator. The relative amount of DC light relative to the slope of the transfer curve at the bias point, which defines the AC component of the output light, is only ½ that of an ordinary MZ, which helps to reduce noise at the receiver.
The device in FIG. 5 has the advantage of requiring only a single RF electrode. A major disadvantage of the LINMOD 30 for some applications is that the CSO components are not suppressed; in fact those components have about the same amplitude as the fundamental frequency components. This device may not be suitable for multi-octave systems, where CSO suppression is required.
FIG. 7 shows a prior art LINMOD 40 using two polarizations within a single conventional MZI 41 made in lithium niobate (LN), as disclosed in L. M. Johnson and H. V. Roussell, “Linearization of an interferometric modulator at microwave frequencies by polarization mixing,” IEEE Photonics Technology Letters, Vol. 2, No. 11, November 1990, pp. 810-811, which is incorporated herein by reference. The LINMOD 40 has a single signal electrode 45, two ground electrodes 43, and a bias electrode structure 44. Light having both orthogonal polarizations is introduced at the input of the device, however, most of the light is in the polarization having the weaker modulation strength.
In the context of this specification, the terms ‘modulation strength’ and ‘modulation efficiency’ are used interchangeably to mean a ratio of the modulation amplitude of a propagation characteristic of light in the waveguide, such as the effective propagation constant of the guided light, to the amplitude of the RF signal in the respective RF electrode that causes the modulation of the propagation characteristic.
The relative strength of modulation for the two polarizations depends primarily on the ratio of the electro-optic tensor coefficients r33 and r13 for each polarization, but also on the overlap integral between the electric field from the applied RF signal and the optical mode profile. For example, in a z-cut LN modulator, the strength of modulation in the TM mode is approximately three times stronger than that found in the TE mode. By launching 96.4% of the light into the TE mode and only 3.6% of the light into the TM mode, the resulting transfer curve has a linear portion near the half power point, as shown in FIG. 8. The device acts as two independent virtual MZIs, one for each polarization, which outputs mix incoherently to linearize the transfer function. Each virtual MZI must be biased independently to the half power (quadrature) point and those bias points must be maintained during operation of the device. The bias electrode structure 44 having three electrodes is used to bias the light in each polarization independently by applying three different voltages to the three electrodes. This results in both vertical and horizontal field components that interact independently with light in the two polarizations.
The conventional LINMOD 40 using two polarizations has several drawbacks. One drawback is that the ratio of optical power in the two polarizations must be carefully maintained to within a percent of target. Another drawback is the need to bias light in two different polarizations. The bias control of two polarizations adds complexity to the system. Any polarization crosstalk after the modulator can seriously degrade CSO suppression. As little as −30 dB polarization crosstalk reduces CSO suppression to only about 20 dB.
U.S. Pat. No. 5,031,235, which is incorporated herein by reference, describes another approach which uses two polarizations. In '235, light from two different light sources that combine incoherently are used instead of light from two orthogonal polarizations. The wavelengths are chosen to be far enough apart such that frequency components from coherent beating are much higher than the bandwidth of the photodetector at the receiver. That '235 device requires two MZ's and hence two RF electrodes instead of one. As before, the frequency response of the RF electrodes must be carefully matched.
An object of the present invention is to provide an improved linearized waveguide modulator utilizing a single RF electrode in at least one modulation section thereof, which obviates at least some of the drawbacks of the prior art modulators.